Dual-mode radiotelephone apparatus for digital or analog modulation

ABSTRACT

A dual-mode radiotelephone capable of operation in analog or digital modes. According to exemplary embodiments, a digital signal processor receives a speech signal to be transmitted in the digital or analog mode and generates In-phase (I) and Quadrature (Q) modulating signals. The I and Q signals are supplied to a quadrature modulator for generating a digitally modulated signal and are supplied to an analog modulator for generating an analog modulated signal.

FIELD OF THE INVENTION

The present invention generally relates to radio transmitters capable ofboth digital and analog modulation for impressing information on thetransmitted signal, and in particular to personal portablecommunications devices such as cellular phones using frequencymodulation or quadrature modulation.

BACKGROUND OF THE INVENTION

In U.S. Pat. No. 5,745,523, filed Oct. 27, 1992, the entirety of whichis incorporated herein by reference, a dual mode radio apparatus havingboth a digital information transmission mode and an analog transmissionmode is described. The modulation in either the digital or analog modeis applied by first computing In-phase (I) and Quadrature (Q) signalsrepresentative of the desired modulated signal vector and applying theI,Q signals through I and Q D/A converters to a quadrature modulator.

U.S. Pat. No. 5,020,076 to Cahill discloses a hybrid modulationapparatus that in a digital modulation mode applies I,Q modulation to aquadrature modulator, and in an analog modulation mode applies an analogfrequency modulation waveform to a phase lock loop to generate a signalwhich is then passed straight through an I,Q modulator which is biasedwith constant I,Q signals.

It would be desirable to generate the analog modulation waveform neededin the Cahill method by using a digital signal processor employed fordigital modulation to compute a sampled digital representation of theanalog modulation waveform, and then converting the digitalrepresentation to the required analog modulation waveform using eitherthe I D/A converter or the Q converter or both.

It would further be desirable for an analog modulation implementation toavoid the Cahill technique of passing the modulated signal through apermanently-biased I,Q modulator, since the Cahill method may sometimesresult in a digital modulation frequency which is undesirable for analogmodulation.

SUMMARY OF THE INVENTION

The present invention is directed toward a transmitter/receiver, such asa radiotelephone, which is capable of operating in two modes. In a firstmode, the transmitter signal is modulated with digital information.Specifically, a digital signal processor computes sampled digitalrepresentations having a real or In-phase waveform (I) and an imaginaryor Quadrature waveform (Q).

After digital-to-analog conversion in respective I and Q D/A converters,a quadrature modulator impresses the I,Q signals on an intermediateradio frequency. The intermediate frequency is subsequently upconvertedto a desired transmission frequency by a local oscillator signal fromthe radiotelephone receiver.

In a second mode, the transmitter signal is modulated with an analogsignal. Specifically, the digital signal processor forms the analogsignal by computing a sampled digital representation of the analogmodulation signal. The sampled digital representation is converted to ananalog waveform using either or both of the I or the Q D/A converter,and applied to an analog modulator to produce an analog modulated radiosignal that may be at a second intermediate frequency. This secondintermediate frequency signal is subsequently converted to the desiredtransmission frequency with the aid of a local oscillator signal fromthe radiotelephone receiver.

In a preferred implementation, both the digital modulation and theanalog modulation are constant envelope modulations that vary only thesignal's phase angle. A preferred method of transferring the desiredangle modulation to the desired transmission frequency is to use avoltage-controlled oscillator to produce a signal at the desiredtransmission frequency and mix the signal with a local oscillator signalfrom the receiver to produce an intermediate frequency signal. Theintermediate frequency signal is phase-compared with either a modulatedor unmodulated reference signal to produce a feedback signal to controlthe oscillator to follow the desired angle modulation waveform. Thefeedback loop bandwidth is furthermore adapted according to themodulation mode selected either to follow digital modulation on thereference signal or not to follow analog modulation applied to thevoltage controlled oscillator.

BRIEF DESCRIPTION OF THE DRAWINGS

The invention will be more readily understood upon reading the followingDetailed Description of the Preferred Embodiments in conjunction withthe accompanying drawings, in which like reference indicia indicate likeelements, and in which:

FIG. 1 is a block diagram of an exemplary dual-mode radiotelephoneaccording to the present invention;

FIG. 2 is a block diagram of an exemplary quadrature modulator suitablefor use in the radiotelephone of FIG. 1;

FIG. 3 is a block diagram showing an exemplary frequency assignmentscheme in the radiotelephone of FIG. 1;

FIG. 4 is a block diagram showing exemplary upconversion circuitrysuitable for use in the radiotelephone of FIG. 1; and

FIG. 5 is a block diagram of an alternative quadrature modulatorsuitable for use in the radiotelephone of FIG. 1.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Referring to FIG. 1, a dual-mode radiotelephone according to anexemplary embodiment of the present invention is shown. The dual-modeapparatus of FIG. 1 is suitable for selectively producing analogmodulation when operating in a first frequency band or digitalmodulation when operating in a second frequency band. For example, thefirst frequency band may be an 800 MHz cellular band in which the analogmodulation conforms to the U.S. AMPS standard while the second frequencyband may be the 1900 MHz “PCS” band recently licensed by the FCC, whichemploys digital modulation according to the GSM standard.

A dual band 800/1900 MHz antenna 10 operates at both frequency bands. Aband-splitting filter 11 separates signals in the two frequency bandsfor application to a diplexing filter 12 for the 800 MHz AMPS bands anda transmission/reception (T/R) switch 13 for the 1900 MHz band in whichtime-duplex is preferably employed. The signals output from diplexingfilter 12 and T/R switch 13 are applied to the inputs of a receiverfront-end 14 for amplification and downconversion to a suitableintermediate frequency or frequencies. The intermediate frequencysignals are filtered and amplified in intermediate frequency processor16 and may be digitized for digital signal processing in DSP 17. Asuitable digitizing method is the logpolar digitizing techniquedescribed in U.S. Pat. No. 5,048,059, the entirety of which isincorporated herein by reference. Front-end 14 includes one or morelocal oscillators to provide mixing frequencies for downconversion, andoutputs one or more local oscillator signals for use in the transmitter15. The local oscillator signal frequency is controlled by frequencysynthesizer 18 which is controllably programmed according to theselected channel frequency. The synthesizer preferably operatesaccording to U.S. Pat. Nos. 5,095,288 and 5,180,993 which are herebyincorporated by reference and provides a fast locking capability frompower up or on channel change to implement the standby power savingfeatures described in U.S. Pat. No. 5,568,513 (Harte, Dent, Croft andSolve), filed May 11, 1993, which is also incorporated herein byreference. The synthesizer 18 may also contain circuits coupled totransmitter 15 for controlling the transmit intermediate frequency.

Transmitter 15 provides transmit power outputs at the analog and digitalfrequency bands. These outputs are fed to antenna 10 via diplexingfilter 12 or T/R switch 13 respectively. T/R switch 13 is controlled byDSP and control processor 17 to allow either transmission or receptionof TDMA signal bursts in an alternating manner known as Time Duplex.Transmitter 15 also receives either analog or digital modulationwaveforms respectively from DSP 17 via the I-waveform connection 19 a orthe Q waveform connection 19 b.

I-waveform modulation connection 19 a preferably supplies balanced Isignals to transmitter 15 via two lines labelled I and {overscore (I)}in FIG. 1. Likewise, the Q-waveform modulation connection 19 bpreferably supplies balanced Q signals to transmitter 15 via Q and Q.These signals are preferably generated by an I,Q modulator such as theone disclosed in U.S. Pat. No. 5,530,722 (Dent, filed Sep. 14, 1994)which is a continuation-in-part of U.S. Pat. No. 5,745,523 and isincorporated herein by reference in its entirety. U.S. Pat. ApplicationNo. 5,530,722 discloses the use of digital converters to generatebalanced I/Q signals from a stream of binary words that numericallyrepresent sampled I/Q waveforms, and high bitrate streams of single-bitsamples and their complements, wherein the desired I or Q value isrepresented by the ratio of binary 1's to binary 0's in a stream. Thisallows the analog waveforms and their inverses to be recovered simply bylow-pass filters, without the use of additional D/A converters.

The balanced, analog I,Q waveforms supplied to transmitter 15 are thenfed to a quadrature modulator composed of an I modulator 15 a and a Qmodulator 15 b to generate respectively a cosine waveform and a sinewaveform at an intermediate transmit frequency TXIF. The I,Q modulatoris preferably performed in transmitter 15, while the cos/sin TXIF signalgeneration may be contained in the transmitter 15 or alternatively inthe receiver section 16, which will be described in more detail belowwith reference to FIG. 3.

As described in U.S. Pat. No. 5,745,523, the balanced I,Q waveformsrepresent digital information modulated in a digital transmission mode,or alternatively represent analog frequency modulation in the analogmode. It will be appreciated that the analog mode may conform to theU.S. AMPS standard or indeed any analog FM cellular standard includingthe British ETACS or Scandinavian NMT systems, all of which employcompanding to compress amplitude variations of speech into a reduceddynamic range.

According to the present invention, an alternative analog modulationmode employing the I,Q modulation lines 19 a and 19 b from DSP section17 can be achieved, wherein the analog modulation is not applied to theI,Q modulator but to a separate analog modulator. FIG. 1 shows anexemplary embodiment wherein the analog modulator is formed by combiningone of the I,Q signals (here the I signal) supplied over an analogmodulation input line 15 c with a phase lock feedback signal fromsynthesizer 18 in a loop filter 15 d. The choice to use the I signal isarbitrary, as the {overscore (I)} signal or one of the Q signals or bothor all could be combined with the phase lock feedback signal in order toachieve analog frequency modulation. The choice of frequency modulationis also arbitrary and amplitude modulation could alternatively be used.The principle of the present invention is illustrated more generally inFIG. 2.

Referring to FIG. 2, it can be seen that DSP 17 includes digital signalprocessor logic 20 for computing either a sampled numerical I,Qrepresentation of a digital modulation waveform or a sampled numericalrepresentation of an analog modulation waveform. The result of thecomputation is applied to at least one of D/A converters 21 a and 21 bwhich preferably operate by converting an input to a high bitratedelta-sigma modulation signal and then low-pass filtering the signal inone of balanced low-pass filters 22 a and 22 b as disclosed in thepatent applications referenced above. The D/A converted, balancedwaveforms are applied to balanced modulators 27 a and 27 b whichmodulate respectively a cosine and a sine waveform at a first transmitfrequency or an intermediate frequency TXIF(D) that is desired for thedigital modulation mode. The digital modulation mode can be, forexample, a GSM mode in which a speech encoder compresses the bitrate andthus the occupied transmission bandwidth for digital speech fortransmission using time-duplex TDMA in the 1900 MHz PCS band; themodulation would be 270-833 kilobits per second for the GSM mode.

If the digitally modulated signal is first generated at an intermediatetransmit frequency by quadrature modulator 25, it is upconverted to thedesired final transmit frequency using an upconverter. For constantamplitude modulation schemes such as GMSK, only the phase of the signalis modulated and must be transferred to the output frequency, which canbe performed by a phase lock loop as will be described-below withreference to FIG. 4.

FIG. 2 further shows that the I,Q signals produced by D/A converters 21a and 21 b can be supplied via select/combine unit 24 to analogmodulator 26 to produce an analog modulated signal at a desiredfrequency TXIF(A) for analog modulation. TXIF(A) can be either thedesired transmission frequency or an intermediate transmit frequencysignal, which is upconverted to the desired transmission frequency aswill be described later with reference to FIG. 4. Select/combine unit 24can be simply a hardwired connection between one of the I,{overscore(I)},Q or {overscore (Q)} signals and modulator 26, but canalternatively combine the I and {overscore (I)} (or Q and {overscore(Q)}) signals in a balanced to unbalanced converter, or perform moresophisticated combining of an I signal representing the coarse part ofan analog modulation waveform with a Q signal representing the errorbetween the coarse representation and the ideal waveform, in order toachieve a reduction in quantizing noise. Another method of combiningseparate I and Q signals to produce a desired modulation would betwo-point frequency modulation of a frequency synthesizer, wherein oneof the I or Q signals is used to frequency modulate a voltage controlledoscillator (VCO) while the other signal is injected into the synthesizerloop controlling the VCO to counteract the loop's tendency to correctthe first signal's modulation of the VCO. Still another method of usingI and Q signals to produce an analog modulated signal would includefrequency modulating the signal with one of I or Q waveforms while usingthe other to determine the amplitude of transmission.

Any of the above methods of selecting one or more of the I,Q signals orusing them in combination to effect analog modulation while bypassingthe quadrature modulator 25 can be implemented in the present invention.

Referring now to FIG. 3, an exemplary frequency plan for a dual-band,dual-mode apparatus according to an embodiment of the invention isshown. To further enhance the present invention, as many components aspossible are reused in both bands or modes in order to achieve the mosteconomical design. In particular, it is desirable to employ adual-frequency antenna (10 of FIG. 1), synthesizer 18, IF amplifier 16etc. for both modes.

One problem in employing a single synthesizer 18 to provide both a firstlocal oscillator frequency and to assist in generating the transmitfrequency is that the duplex spacing (the frequency difference betweentransmit-receive frequency pairs) may not be the same in the digital andanalog modulation mode, or in the 800 MHz cellular band compared to the1900 MHz PCS band. The synthesizer 18 produces a local oscillatorfrequency (L01) that is higher (for example) than the receive channelfrequency Frx by the desired first intermediate frequency IF1. To usethe same value of L01 to produce the transmit frequency when operatingin the same band (800 MHz or 1900 MHz) the apparatus must include meansto offset L01 by an amount TXIF(A or D) to produce the desired transmitchannel frequency Ftx. It can readily be seen that

Ftx=L 01−TXIF

Frx=L 01−IF 1

Therefore DUPLEX SPACING=Frx−Ftx=TXIF−IF1 or alternativelyTXIF=IF1+DUPLEX SPACING.

If substantially the same frequency IF1 is to be used in both modes, butthe duplex spacing is different, then TXIF(A) in one mode will not bethe same as TXIF(D) in the other mode. This hinders the use of the sameI,Q modulator for modulation in both modes, as it may be difficult toproduce an I,Q modulator that performs well at both TXIF(A) and TXIF(D).The actual duplex spacings are 45 MHz in the 800 MHz band and 80 MHz inthe 1900 MHz band. Therefore TXIF(A) and TXIF(D) will differ byapproximately 80−45=35 MHz if the same IF1 is to be used. It would ofcourse be possible to configure a design which uses the same TXIF inboth bands and modes, by instead allowing different IF1s. However, thishinders the use of common components in the receive path and increasescost and complexity. The preferred arrangement uses values of IF1 thatare the same or nearly the same in both modes, thus allowing TXIF(A) andTXIF(D) to be separately adapted to produce the desired duplex spacings.This is facilitated by the invention which provides an economic way toperform analog modulation using a TXIF(A) that is different from theTXIF(D) employed by the quadrature modulator 25 for digital modulation.

FIG. 3 shows a preferred frequency plan based on a 13 MHz referenceoscillator. 13 MHz is the basis for deriving the GSM bitrate (13 MHz/48)and the GSM channel spacing (13 MHz/65) in the 1900 MHz band. To obtain30 KHz channel steps in the 800 MHz band, synthesizer 18 needs areference frequency that is a multiple of 30 KHz, and preferably amultiple of 8×30 KHz in order to provide fast channel changing time andlow phase noise by use of the fractional-N technique of incorporatedU.S. Pat. No. 5,180,993. Neither 30 KHz nor 240 KHz divides into 13 MHz,but this problem is solved by multiplying the frequency from referenceoscillator 30 by six in frequency multiplier 31 to obtain 78 MHz, whichis 325 times 240 KHz. 78 MHz is also a convenient second localoscillator frequency for mixing signals at the chosen first IF frequencyof 72 MHz down to a second intermediate frequency of 6 MHz in secondmixer 33. The 6 MHz signal is then further amplified and filtered insecond IF 34 to produce an output for digitizing and processing in DSP17. The choice of second intermediate filter equal to 6 MHz allows theuse of existing off-the-shelf filters available at that frequency, butit will be appreciated that other suitable frequencies may be chosen.

In the time-duplex TDMA digital mode (GSM at 1900 MHz), the duplexspacing is 80 MHz, which requires a TXIF(D) equal to IF1 (=72 MHz) plus80 MHz, or 152 MHz. By further frequency doubling the 2nd localoscillator signal L02 from 78 MHz to 156 MHz, a useable TXIF(D) isobtained. The 4 MHz discrepancy between 152 and 156 MHz can beaccommodated because transmission and reception do not take place at thesame time. Therefore, using a fast switching synthesizer 18, the valueof L01 may be changed 4 MHz between reception and transmission. Tofacilitate this, the phase comparator reference frequency used bysynthesizer 18 in the digital mode is 1 MHz, derived by dividing the 78MHz signal by 78 in reference divider 40. Fractional-N synthesizercircuit 41 then further interpolates by a factor of 5 to obtain thedesired 200 KHz steps of GSM. In the analog mode, 30 KHz steps areobtained by first dividing 78 MHz by 325 in reference divider 40, theninterpolating by a factor of 8 using the fractional-N synthesizer 41.Synthesizer 18 is appropriately programmed for these different modes andchannel frequencies by control signals sent from control processor ofDSP 17.

The frequency multipliers used to produce 78 MHz and 156 MHz can employphase lock loops that divide an oscillator running at the desiredfrequency by 6 or 12 to produce a 13 MHz signal which is compared withthe 13 MHz signal from oscillator 30 to produce a feedback controlsignal to control the oscillator to the desired multiple. For example, a156 MHz signal can be produced in this manner and a 78 MHz signal can bederived from an intermediate divide-by-2 output of the loop divide-by-12circuit.

The 156 MHz signal is modulated by digital information in the digitalmode by I,Q modulator 25, which includes quadrature network 28 togenerate cosine and sine waveforms. The modulated signal at TXIF(D)(=156 MHz) is then upconverted to the desired 1900 MHz band in thefollowing manner applicable to constant envelope modulations: VoltageControlled Oscillator 54 operates at the desired output frequency todrive transmit power amplifier 55. The output signal from oscillator 54is mixed with the receive local oscillator signal L01 which has beensidestepped 4 MHz as described above during the transmit burst. Theresulting signal after low pass filtering in filter 51 is at TXIF(D) andis phase compared in comparator 50 with the I,Q modulated signal atTXIF(D) from quadrature modulator 25. Comparator 50 generates a phaseerror signal which, after loop filtering using an integrator in filter53 produces a control signal coupled to VCO 54 that controls its phaseto follow the modulation imposed on TXIF(D) by modulator 25. In thisway, the phase modulation is transferred to the transmit frequency.

In analog mode however, IF1 is chosen to be 72.06 MHz. This choice of anIF1 for analog mode that is nearly the same as for digital mode;however, the same IF1 of 72 MHz could have been chosen, or alternatively71.94 MHz. Small differences such as 60 KHz in IF1 between analog anddigital mode do not prejudice the use of the same 6 MHz IF amplifier 34as its filter bandwidth is sufficiently broad to encompass 5.94 MHz aswell as 6 MHz or 6.06 MHz.

The duplex spacing of 45 MHz in the 800 MHz band leads to a TXIF(A)equal to 72.06+45=117.06 MHz. This is a multiple (×1951) of 60 KHz andmay easily be produced using auxiliary synthesizer 43, sharing referencedivider 40 with main synthesizer 41. The auxiliary synthesizer mayfurther divide the 240 KHz reference signal by 4 to obtain the desired60 KHz reference signal, at which frequency it is phase-compared withthe signal TXIF(A) divided by 1951.

An oscillator 64 running at the desired transmit frequency in the 800MHz cellular band drives power amplifier 65 and is also coupled to mixer62 where it is mixed against the receive local oscillator L01 operatingat the appropriate frequency for reception in the 800 MHz band. Theresulting signal at 117.06 MHz is applied, after low pass filtering infilter 61, to auxiliary synthesizer 43 where it is divided by 1951 to 60KHz and phase compared to the 60 KHz reference signal to produce a phaseerror signal.

The phase error signal is filtered and integrated in loop filter 63 andthen added to the selected I or Q or combined signal, the desired analogfrequency modulating waveform to produce a control signal for oscillator64. The control signal limits the oscillator 64 to the desired transmitchannel frequency and modulates the oscillator 64 with the desiredfrequency modulation waveform.

It should be noted that while separate mixers 52,62, low-pass filters51,61 and power amplifiers 55,65 are illustrated in FIG. 3, any or eachpair can be combined to reduce complexity. For example, mixers 52 and 62can be the same mixer, the selection of which is based on which ofoscillators 54,64 was enabled to drive it. Had the value of IF1 for theanalog mode been chosen instead to be 72 MHz, the value of TXIF(A) wouldbe 117 MHz, which is 1950×60 KHz or 975×120 KHz. This would allow thechoice of a 120 KHz phase-comparison frequency in the auxiliarysynthesizer. Alternatively, the 117 MHz is nine times the 13 MHzreference crystal frequency, which can be generated using the sametechnique as for the 156 MHz TXIF(D). This would be a good choice if aquadrature modulator operating at both 117 MHz and 156 MHz used,together with I,Q modulation in analog mode. However, producing widebandI,Q modulators is difficult and costly, and therefore the exemplaryembodiments of the present invention teach the use of one or more of theI,Q D/A converted signals as a conventional analog modulation signalapplied to the auxiliary synthesizer loop. When the synthesizer loop isused to control a phase or frequency modulated signal, it is desirableto have a large division factor (1950) in the loop in order to reducethe amount of modulation reaching the phase comparator at 60 or 120 KHz.This is because a large phase error signal due to the modulation cancause distortion of the modulation frequency response or non-lineardistortions due to non-linearities of the phase detector.

Thus, the exemplary frequency plan of FIG. 3 includes a range of optionsfor the values of IF1 and TXIF(A) used in the analog modulation mode.The choice of IF1, in the analog mode to be 72.06 or 71.94 MHz can beuseful also in the digital mode. According to the digital GSM standard,the cellular network radiates a special signal called the FrequencyCorrection Burst or Frequency Correction Channel (FCH) as part of theBroadcast Control Channel (BCCH). The FCH is an unmodulated TDMA burst.More precisely, the burst is modulated by an all-1's or all-0's pattern,which produces a CW carrier that is offset by ¼ of the bitrate, due tothe characteristics of Gaussian Minimum Shift Keying modulation (GSMK).Since the bitrate is 13 MHz/48 (270.833 KB/s), the frequency is offsetby +67.708 KHz. This is translated to the first intermediate frequencywith a sign change if the receive local oscillator frequency L01 isgreater than the receive frequency Frx. The FCH burst may be readilydetected by using a narrow bandpass filter such as filter 38 of FIG. 3,centered on the offset frequency. Since the bandwidth of filter 38 ischosen to be approximately ±15 KHz for analog mode, it is approximatelyof the correct width to detect the FCH burst in digital mode, despitethe selection of filter 38 to be centered on 72.06 with an FCH frequencyof 72.067708 or 71.94 with an FCH of 71.932292 MHz. The error of +7.708KHz is within the ±15 KHz bandwidth of the filter and so passes throughto DSP 17 for detection. DSP 17 may digitally correct for the 7.708 KHzof error and may further reduce the bandwidth prior to detection.Detection can be performed by monitoring the signal energy out of thenarrowband filtering to determine if it increases at the predefined FCHburst interval repetition rate compared to energy outside the FCH burstintervals. All possible interval timings can be explored. The timingwhich exhibits the greatest increase in narrowband energy is used toestablish coarse TDMA network timing, and then other signal and messagecontent are searched for in the BCCH signal.

FIG. 4 shows more detail of the upconversion process from intermediatetransmit frequencies TXIF(A) or TXIF(D) to the final transmit frequencyFtx in either the 800 MHz or 1900 MHz bands.

Quadrature modulator 25 (FIG. 2), composed of balanced I-modulator 27 aand Q-modulator 27 b, is driven by cosine and sine outputs fromQuadrature VCO 84 through buffers 86,87. A buffered output throughbuffer 85 is also coupled to divide by 12 circuit 81,83 having anintermediate divide by 2 output at 78 MHz coupled through buffer 82 toIF amplifier 16. This 78 MHz output is used in a second downconvertingmixer in IF amplifier 16, which is preferably of the image rejectiontype. An image rejection mixer also requires cosine and sine injectionwaveforms at 78 MHz, and can be derived easily from the 156 MHz signalby producing a 78 MHz divide-by-2 output delayed by one half cycle ofthe 156 MHz waveform, as an undelayed output from buffer 82. The extracomponents for this embodiment are not shown in FIG. 4, but suitablecomponents will be readily apparent to those skilled in the art.

The output of divide by 6 circuit 81 from 78 MHz is at 13 KHz, the sameas the reference crystal oscillator signal frequency, with which it iscompared in phase comparator 80 having a bipolar current-mirror output.A phase error signal output by comparator 80 is produced in the form ofa current proportional to the phase error, which can be low-passfiltered and integrated using purely passive loop filter components 88.The filtered and integrated error signal is fed back to control QVCO 84to the desired 156 MHz frequency. The loop is preferably a second orderservo having a fast lock time that enables rapid power-up and power-downin order to reduce battery consumption by activating the entire QVCO andits control loop only during digital TDMA receive timeslots and digitaltransmit timeslots, or during analog receive periods. The duty factorduring reception of the analog control channel, to which the apparatuslistens on standby, may be minimized using known battery savingtechniques.

The outputs of modulators 27 a, 27 b are summed through low-pass filter51 and applied to phase detector 50, where the GMSK modulated signal iscompared with a signal derived from transmit power amplifier 55 bydownconversion against the receive local oscillator in mixers 52,62.

FIG. 4 illustrates the previously mentioned inventive feature ofcombining certain parts common to both frequency bands. Mixers 52,62have been combined into a single mixer with selection of the signalsource by the enable inputs of input buffers 521,621. When transmissionin the 1900 MHz band is required, buffer 521 is enabled by a signalEN1900 from DSP 17 to allow a sample of the 1900 MHz transmit signaltaken by coupler 552 to pass to the mixer 522. Alternatively the signalEN800 is activated to pass a sample of the 800 MHz signal to mixer 522.The local oscillator from dual-band front end 14 is selected at itssource to deliver a signal appropriate to the selected band in order toconvert the selected transmit signal frequency to the desired transmitIF TXIF(A) to TXIF(D). The converted signal output is low-pass filteredin combined filter 51,61 and fed by dual-output buffer 81 to both phasedetector 50 and auxiliary synthesizer circuit 43 comprising divider 90,reference divider 91 and phase detector 92.

When a 1900 MHz transmit operation is selected, quadrature modulator 25and phase detector 50 are powered up. Phase detector 50 compares thephase of the modulated TXIF(D) at 156 MHz with the output from buffer 89to produce a current signal proportional to the phase error. The currentis filtered and integrated using only passive filter components 53.Optionally, the phase lock time from power up can be minimized byrecording previous voltages on loop filter capacitor 531 at the end of aTDMA burst transmission in a look-up table, versus channel frequency,within DSP/Control processor 17.

The previously digitized and stored voltage value is then recalled, D/Aconverted, and applied to principal integrator capacitor 531 toprecharge the capacitor to approximately the correct voltage just beforetransmission of a TDMA burst on the same previous frequency, thusreducing relock time. When the loop is locked, the closed-loop bandwidthprovided by loop filter 53 is preferably wide enough to cause the phaseof the transmit signal sample to be controlled to follow the phase ofthe GMSK signal from quadrature modulator 25.

When transmission in the 800 MHz band is selected, the output frombuffer 89 is divided by 1951 in programmed divider 90 and compared witha 60 KHz reference obtained by further dividing the 240 KHz output ofdivider 40 by four in divider 91. The comparison of the two 60 KHzsignals in phase detector 92 produces an error current signal whose meanvalue is proportional to the phase error which can be low-pass filteredand integrated using passive loop-filter 63 to obtain a control signalfor 800 MHz transmit VCO 64. Passive loop filter 63 includes components634, 635, 633, 632 and 631 designed to filter a balanced I signalI′,{overscore (I)}′ from DSP 17 to produce the balanced I,{overscore(I)} drive signals from modulator 27 a as well as to filter common modecomponents of I′,{overscore (I)} with a different filter characteristicfor injection into the loop via capacitors 634,635. The arrangement ofthis filter illustrates the option of using both I and {overscore (I)}signals to produce analog modulation. In this case, I′ and {overscore(I)}′ signals from DSP 17 are not complementary signals, but areselected to be the same or to have a sum or mean value representing thedesired analog modulation. The phase lock loop filter will pass themodulation signals to VCO 64 causing phase or frequency modulation. Thefeedback signal from phase detector 92 will tend to counteract thismodulation, but this tendency is reduced by divider 90. Nevertheless,the modulation at the lowest modulation frequencies will be partiallycounteracted, perhaps necessitating boosting the low modulationfrequencies prior to D/A conversion in DSP 17. Such a boost combinedwith other filtering operations in the digital domain in DSP 17 aredesigned to achieve the overall modulation frequency response desired inanalog FM mode, including pre-emphasis.

In one implementation PA 65, oscillator 64 and mixer 62 are constructedas a single unit using a Gallium Arsenide (GaAs) integrated circuit,while 55, 54 and 52 form a second GaAs circuit. Both GaAs circuits mayalso be combined into a single dual-band GaAs integrated circuit.

In the digital mode, loop filter 53 is preferably designed to obtain aclosed loop transfer function broad enough to follow the digitalmodulation. In the analog mode, loop filter 63 is preferably narrow, soas to prevent the loop from attempting to correct for the impressedanalog modulation. Specifically, by tailoring the characteristics withthe loop integrating filters 53,63, the desired angle modulation istransferred to the oscillator at the output frequency while suppressingnoise at other frequencies and in particular in the receive frequencyband. A first loop filter characteristic is employed for AMPSmodulation, which has desired modulation components only up to about 10KHz, while a second loop filter characteristic is employed for a GSM270.833 KB/S GMSK waveform that has desired components up to 150 KHz.The wider loop filter characteristic in the GSM case allows rapidacquisition of phase lock at the beginning of the TDMA burst such thatthe VCO accurately tracks the desired angle waveform only a few tens ofmicroseconds after the circuit is enabled. The phase lock circuit maythus be powered down to save power during the receive part of the TDMAframe and powered up just prior to the transmit part of the frame.

The invention can include the use of an analog-to-digital converter tomeasure the loop filter integrator voltage at the end of a burst and torecord the value numerically in a microprocessor memory against thechannel frequency. When the same channel frequency is selected later,the voltage value is recalled and applied to a D/A converter toprecharge the loop integrator, e.g., during the receive or idle part ofthe TDMA frame. Just prior to the transmit part of the frame, the D/Aconnection to the loop integrator capacitor is open-circuited(tri-stated) and the phase error current is used to finely adjust theoscillator control voltage under closed loop control. Aftertransmission, the find-adjusted voltage may be read to overwrite theprevious value in memory to provide continuous recalibration. Thisfeature can be used to speed up phase lock acquisition and can also beused in the analog FM mode by recording the loop voltage againstfrequency eery time the transmitter is operated at any frequency.

An alternative embodiment incorporating the aforementioned Cahilldisclosure is shown in FIG. 5. Control processor/DSP 17 in this caseproduces I′,{overscore (I)}′ and Q′,{overscore (Q)}′ signals which arefiltered using balanced filter 63 a to obtain a first balanced, filteredsignal I,{overscore (I)} to drive modulator 27 a, a second balancedfiltered signal Q,{overscore (Q)} to drive modulator 27 b, and a commonmode filtered signal I″ to inject into combined loop filter 53,63. Inthe digital mode, the common mode signal is suppressed by causing I′ and{overscore (I)}′ to be generated as complementary signals. In the analogmode, the balanced mode signal is suppressed by generating I′ and{overscore (I)}′ to be non-inverted signals. This results in modulator27 a having nominally no output. The output in the analog mode is causedby supplying constant signals Q,{overscore (Q)}′ at their maximumcomplementary levels to allow the QVCO signal at 117 MHz to pass throughQ-modulator 27 b at the same time the QVCO loop receives an injection ofanalog modulation from the common mode I′,{overscore (I)}′ signal. Inaddition, reference divider 40 and auxiliary divider 43 are programmedto different values depending on whether the digital or analogmodulation mode is selected.

An improvement over Cahill may be achieved by programming DSP 17 togenerate both analog and I,Q modulation substantially simultaneously. Inthis mode, high-frequency components of the desired analog modulationare generated in the common mode part of the I′,{overscore (I)}′ signalsand modulate frequency or phase angle inside the QVCO loop. Lowfrequency components of the desired angle modulation which wouldotherwise be counteracted by the loop's feedback action are appliedoutside the loop by generating the balanced part of the I′,{overscore(I)}′ and Q′,{overscore (Q)}′ signals to be proportional to the cosineand sine respectively of the desired low-frequency phase modulationsignal. In this way, phase modulation down to zero frequency can beobtained. Optionally Q and {overscore (Q)}′ can also be fed to loopfilter (53,63) and the common-mode part of Q,{overscore (Q)}′ can beused together with the common-mode part of I′,{overscore (I)}′ toprovide enhanced accuracy for frequency-modulation VCO 84. In general,the I signal can be generated as a delta-modulation representation ofhalf the sum of a desired balanced (odd mode) and unbalanced (even mode)waveform, while {overscore (I)}′ is a delta-modulation representation ofhalf the difference. Likewise, Q and {overscore (Q)}′ represent half thesum and half the difference of desired even and odd waveforms.

The selection of division ratios has been previously described. Theselection of low division ratios 6 and 12 respectively results in a highloop bandwidth in the digital mode for controlling the frequency of QVCO84 to 156 MHz. The selection of high division ratios 1300 and 1951 inthe analog mode result in a low loop bandwidth for controlling QVCO 84to 117.06 MHz. By suitably selecting division ratios and designing loopfilters 53,63 according to well known techniques, different loopbandwidths may be produced even at the same intermediate transmitfrequency (e.g., 117 MHz) in the case that analog and digital modulationwas required in the same frequency band, for example for the purpose ofimplementing the dual-mode phone of U.S application Ser. No. 07/967,027.In that case, the modulation in digital mode is not a pure phasemodulation but also includes amplitude modulation, so that upconverter100 is a linear upconverter to the final transmit frequency, and poweramplifier 551 is a linear PA.

The power amplifier requirements for implementing constant amplitudemodulations as used in GSM and AMPS are easier than for the varyingamplitude modulations used in D-AMPS, therefore it is preferable from atransmitter point of view to combine GSM and AMPS standards into adual-mode phone. On the other hand, AMPS and GSM use different channelbandwidths and spacings of 30 KHz and 200 KHz respectively. From areceiver point of view it is, according to the prior art, preferable touse the same bandwidth in both modes. The inventive architecture of adual mode phone disclosed here economically achieves both low cost,dual-band or single band, digital and analog modulation of thetransmitter while accommodating both wide and narrow band filtering forreception of GSM, AMPS or D-AMPS respectively.

A further aspect of the present invention provides for facilitating theconstruction of radio transmitter-receivers, such as digital cellularphones or personal wireless communicators, that operate according to twodifferent standards using non-integrally related bandwidths or bitrates.

In an exemplary implementation, a radio receiver for GSM and AMPSsignals includes a reference clock at a frequency of 39 MHz, which isdivided by 144 to produce a first sampling rate of 270.833 KS/S, oralternatively divided by 150 to produce a second sampling rate of 260KS/S. When receiving GSM (digital) signals, the received signals aredigitized by any suitable means (Cartesian or Logpolar) using the firstsample rate, and when receiving AMPS (analog) radio signals the receivedsignals are digitized using the second sample rate. The analog signalstream sampled at the second (260 KS/S) rate is then digitally filteredto narrow the receiver pass bandwidth to a value adapted to the AMPSmode, and simultaneously the sampling rate is reduced by downsampling to80 KHz, being a convenient multiple of a 10 KB/S signalling rate used inAMPS and of a standard 8 KS/S speech processing rate for processingsampled and digitized (PCM) speech.

Downsampling in a digital filter is well known when the output samplerate is an integral submultiple of the input sample rate. In certainapplications as exemplified above, it can be desirable to produce anoutput sample rate that is not an integral submultiple of the inputrate, i.e. 80:260 or 4:13. The present invention provides a means ingeneral to compute N output samples for every M input samples in adigital downsampling filter.

According to an exemplary embodiment, the inventive method comprisesusing N digital filters such as FIR filters each having an associatedset of filter coefficients adapted to compute one output sample per Minput samples. The coefficient are chosen such that the output samplescomputed by successive filters represent a filtered signal value at Nsuccessive time intervals equispaced over each period of M inputsamples. In an exemplary application, a 260 KS/S input sample rate isapplied to four filters, each of which produces an output sample at a 20KHz rate. The four 20 KHz streams may then be multiplexed to produce an80 KS/S stream which is further processed to extract 10 KB/S Manchestercoded and frequency modulated signalling data or 8 KS/S PCM speech whichis then D/A converted using a PCM CODEC circuit to produce an analogspeech waveform that is fed to an earpiece.

Many variations to the illustrative embodiments disclosed above will bereadily apparent to a person skilled in the art without departing fromthe spirit and scope of the invention as defined by the following claimsand their legal equivalents.

What is claimed is:
 1. A radio transmitter/receiver for selectivelytransmitting digitally modulated signals in a digital mode or analogmodulated signals in an analog mode comprising: digital signalprocessing means having an input for receiving an information signal, afirst output for supplying an In-phase modulating signal I and a secondoutput for supplying a Quadrature modulating signal Q; quadraturemodulation means coupled to the first and second outputs for digitallymodulating a carrier frequency in the digital mode to produce adigitally modulated signal; and analog modulation means coupled to thefirst and second outputs for generating an analog modulation of acarrier frequency in the analog mode to produce an analog modulatedsignal, wherein said In-phase modulating signal I and said Quadraturemodulating signal Q bypass said quadrature modulation means in order toproduce said analog modulated signal.
 2. The radio transmitter/receiverof claim 1, wherein the digital signal processing means includes logiccircuitry for computing a sampled numerical representation of theinformation signal, D/A conversion means for converting the samplednumerical representation into the In-phase modulating signal I and theQuadrature modulating signal Q.
 3. The radio transmitter/receiver ofclaim 1, wherein the digitally modulated signal is at a first frequency,the analog modulated signal is at a second frequency, and thetransmitter further comprises upconversion means for converting thedigitally modulated signal to a third frequency and converting theanalog modulated signal to a fourth frequency.
 4. The radiotransmitter/receiver of claim 3, wherein the upconversion means includesdigital and analog loop integrating filters which control digital andanalog voltage controlled oscillators, respectively, to convert thedigital modulated signal to the third frequency and the analog modulatedsignal to the fourth frequency, while suppressing noise at otherfrequencies.
 5. The radio transmitter/receiver of claim 4, wherein thedigital loop integrating filter has a wider transfer function than theanalog loop integrating filter.
 6. The radio transmitter/receiver ofclaim 1, wherein the transmitter operates according to the GSM standardin the digital mode, and according to the AMPS standard in the analogmode.
 7. The radio transmitter/receiver of claim 1, wherein the digitalsignal processing means generates the In-phase modulating signal I andthe Quadrature modulating signal Q substantially simultaneously.
 8. Theradio transmitter/receiver of claim 4, wherein the upconversion meansfurther includes means for measuring and storing a loop filterintegrator voltage and a corresponding frequency after each receivedtransmission burst, and for precharging one of the loop integratingfilters using a previously stored loop filter integrator voltage priorto transmission on a corresponding frequency.
 9. The radiotransmitter/receiver of claim 1, wherein digitally modulated signals oranalog modulated signals are received, and received digitally modulatedsignals are digitized at a first sampling rate and received analogmodulated signals are digitized at a second sampling rate, and digitizedanalog signals is narrowed, and the second sampling rate is reduced toan output sampling rate.
 10. The radio transmitter/receiver of claim 9,wherein the output sampling rate is not an integral submultiple of thesecond sampling rate.
 11. The radio transmitter/receiver of claim 9,wherein there are N digital filters, each having an associated set offilter coefficients for computing one output sample for every M inputsamples.
 12. The radio transmitter/receiver of claim 11, wherein thefilter coefficients are chosen such that the output samples computed bysuccessive filters represent a filtered signal value at N successivetime intervals equally spaced over each period of M input samples. 13.The radio transmitter/receiver of claim 1, wherein said analogmodulation means includes means for selecting or combining said firstand second outputs in order to generate a frequency or amplitudemodulating signal.
 14. A radio transmitter/receiver for selectivelytransmitting a bandwidth-compressed digital speech signal or anamplitude-compressed analog speech signal comprising: analog-to-digitalconversion means having an input for receiving an analog speech signaland an output for producing a sample stream of numerical samplesrepresentative of the analog speech signal; digital signal processingmeans having an input for receiving the sample stream and one or moreoutputs for supplying an In-phase modulating signal I and a Quadraturemodulating signal Q, the digital signal processing means converting thesample stream into a bandwidth-compressed and coded digital speechsignal and an I and a Q sample stream representative of a digital vectormodulation of the bandwidth-compressed digital speech signal or anumerical stream on at least one of the outputs representative of anamplitude-companded version of the analog speech signal;digital-to-analog conversion means for converting the I and Q samplestreams to associated I and Q analog modulating waveforms; quadraturemodulator means coupled to the digital-to-analog conversion means forvector modulating a carrier frequency with the I and Q analog modulatingwaveforms for transmitting the bandwidth-compressed digital speechsignal; and analog modulation means coupled to the digital-to-analogconversion means for producing an analog modulation of a carrierfrequency using at least one of the I or Q analog modulating waveformsfor transmitting the amplitude-compressed analog speech signal.
 15. Theradio transmitter/receiver of claim 14, wherein the bandwidth-compresseddigital speech signal is at a first frequency, the amplitude-compressedanalog speech signal is at a second frequency, and the transmitterfurther comprises upconversion means for converting the digital speechsignal to a third frequency and converting the analog speech signal to afourth frequency.
 16. The radio transmitter/receiver of claim 15,wherein the upconversion means includes digital and analog loopintegrating filters which control digital and analog voltage controlledoscillators, respectively, to convert the digital speech signal to thethird frequency and the analog speech signal to the fourth frequency,while suppressing noise at other frequencies.
 17. The radiotransmitter/receiver of claim 16, wherein the digital loop integratingfilter has a wider transfer function than the analog loop integratingfilter.
 18. The radio transmitter/receiver of claim 16, wherein theupconversion means further includes means for measuring and storing aloop filter integrator voltage and a corresponding frequency after eachreceived transmission burst, and for precharging one of the loopintegrating filters using a previously stored loop filter integratorvoltage prior to transmission on a corresponding frequency.
 19. Theradio transmitter/receiver of claim 14, wherein the transmitter operatesaccording to the GSM standard for transmitting the digital speechsignal, and according to the AMPS standard for transmitting the analogspeech signal.
 20. The radio transmitter/receiver of claim 14, whereinthe digital signal processing means generates the In-phase modulatingsignal I and the Quadrature modulating signal Q substantiallysimultaneously.
 21. The radio transmitter/receiver of claim 14, whereindigitally modulated signals or analog modulated signals are received,and received digitally modulated signals are digitized at a firstsampling rate and received analog modulated signals are digitized at asecond sampling rate, and the bandwidth of digitized analog signals isnarrowed, and the second sampling rate is reduced to an output samplingrate.
 22. The radio transmitter/receiver of claim 21, wherein the outputsampling rate is not a integral submultiple of the second sampling rate.23. The radio transmitter/receiver of claim 22, wherein there are Ndigital filters, each having an associated set of filter coefficientsfor computing one output sample for every M input samples.
 24. The radiotransmitter/receiver of claim 23, wherein the filter coefficients arechosen such that the output samples computed by successive filtersrepresent a filtered signal value at N successive time intervals equallyspaced over each period of M input samples.
 25. An apparatus foralternatively modulating a carrier signal with an analog signal or adigital signal, comprising: digital signal processing means forgenerating a first pair of delta-modulation bitstreams representative ofeither a balanced In-phase component of a digitally modulated signal oran unbalanced high-frequency component of an analog frequency modulatedsignal plus an In-phase balanced low-frequency component, and a secondpair of delta-modulation bitstreams representative of either a balancedQuadrature component of a digitally modulated signal or a balancedQuadrature low-frequency component of an analog modulation waveform;Quadrature modulator means, responsive to the balanced In-phase andQuadrature signals, for Quadrature-modulating the carrier signal; andfrequency modulation means responsive to the unbalanced high-frequencycomponent for frequency modulating the carrier signal, wherein thecarrier signal is either quadrature modulated with a digital modulationsignal or frequency or phase modulated with both a low-frequency and ahigh-frequency component of an analog modulation signal.
 26. Theapparatus of claim 25, wherein the digital signal processing meansgenerates the In-phase and Quadrature signals substantiallysimultaneously.
 27. A radio transmitter/receiver for selectivelytransmitting digitally modulated signals in a digital mode or analogmodulated signals in an analog mode comprising: a digital signalprocessor having an input for receiving an information signal, a firstoutput for supplying an In-phase modulating signal I and a second outputfor supplying a Quadrature modulating signal Q; a quadrature modulatorcoupled to the first and second outputs for digitally modulating acarrier frequency in the digital mode to produce a digitally modulatedsignal; a selecting unit coupled to the first and second outputs forselecting said In-phase modulating signal I and said Quadraturemodulating signal Q and generating a modulating waveform, or a combiningunit coupled to the first and second outputs for combining said In-phasemodulating signal I and said Quadrature modulating signal Q andgenerating a modulating waveform; and an analog modulator for receivingsaid modulating waveform and producing an analog modulated signal. 28.The radio transmitter/receiver of claim 27, wherein said selecting unitcomprises a hardwire connection between said first and second outputsand said analog modulator.
 29. The radio transmitter/receiver of claim27, wherein said combining unit includes a voltage controlledoscillator; and wherein one of said In-phase modulating signal I or saidQuadrature modulating signal Q is used to frequency modulate the voltagecontrolled oscillator while the other signal is injected into asynthesizer loop controlling the voltage controlled oscillator.
 30. Amethod for selectively transmitting digitally modulated signals in adigital mode or analog modulated signals in an analog mode comprisingthe steps of: phase receiving an information signal in a digital signalprocessor and generating an In-phase modulating signal I and aQuadrature modulating signal Q; determining whether a digital mode or ananalog mode has been selected; digitally modulating, in a quadraturemodulator, a carrier frequency to produce a digitally modulated signalif said digital mode has been selected; selecting, if said analog modehas been selected, an In-phase modulating signal I or a Quadraturemodulating signal Q, or combining an In-phase modulating signal I and aQuadrature modulating signal Q, to produce a modulating waveform; andgenerating, in an analog modulator, an analog modulation of a carrierfrequency to produce an analog modulated signal using said modulatingwaveform if said analog mode has been selected.
 31. The method of claim30, further comprising the step of: upconverting said digitallymodulated signal, if said digital mode has been selected, from a firstfrequency to a second frequency; and upconverting said analog modulatedsignal, if said analog mode has been selected, from a third frequency toa fourth frequency.